Wireless power apparatus and methods

ABSTRACT

Wireless energy transfer system. Antennas are maintained at resonance with High Q. Techniques of maintaining the high-Q resonance matching are disclosed.

RELATED APPLICATIONS

This application is related to co-owned and co-pending U.S. patentapplication Ser. No. 11/408,793 Filed Apr. 21, 2006 and entitled “Methodand System for Powering an Electronic Device Via a Wireless Link”, andU.S. patent application Ser. No. 11/654,883 filed Jan. 17, 2007 entitled“Method and Apparatus for Delivering Energy to an Electrical orElectronic Device Via a Wireless Link”, each of the foregoingincorporated herein by reference in its entirety.

This application claims priority from provisional application No.60/904,628, filed Mar. 2, 2007; the disclosure of which is herewithincorporated herein by reference.

COPYRIGHT

A portion of the disclosure of this patent document contains materialthat is subject to copyright protection. The copyright owner has noobjection to the facsimile reproduction by anyone of the patent documentor the patent disclosure, as it appears in the Patent and TrademarkOffice patent files or records, but otherwise reserves all copyrightrights whatsoever.

All figures, tables, and Exhibits are Copyright© 2006, 2007 ThirdOption, LLC. All rights reserved.

BACKGROUND

Delivery of power to portable devices often uses wires of various typesto carry out the power delivery. Devices such as cell phones, portablecomputers, or any other device that can operate from stored power suchas a battery, all require and use such a source of power to operate thedevice and/or charge the battery.

SUMMARY

Techniques of wireless power transfer are disclosed herein.

DETAILED DESCRIPTION

Embodiments describe the use of wireless power transfer to a receivingsource.

As used herein, the terms “wireless power” and “wireless energy” includewithout limitation any form of power or energy, including thatassociated with electric fields, magnetic fields, electromagneticenergy, or otherwise, that is transmitted between one point, area,location or device and another without having to use a wire lineconnection.

An embodiment discloses a wireless powering and charging system. Anembodiment describes using a transmitter of a size allowing it to beembedded into another item, e.g., a desk or a shelf, or plugged into awall, or embedded into another structure or surface such as a wall,floor, door, etc. A receiver is associated with a small mobile unit orclient device carried by the user, or mounted on a portable device,vehicle, or with a stationary device such as a lamp, toaster,flat-screen TV on a wall, computer or computerized device, PDA, personalmedia device, etc. When the receiver is in range of the transmitter,power is delivered to the mobile unit.

In one embodiment, a wireless powering-charging system is disclosed,based on a transmitter that sends a substantially unmodulated signal orbeacon (e.g., the carrier only). A receiver may be tuned to extractenergy from the radiated field of the transmitter. The receiver powersan electronic device or charges a battery.

Other embodiments may use beacons that are slightly modulated.

Multiple receivers may be used. Multiple transmitters may be used totransmit to one or multiple receivers.

The antenna used by this system allows an efficient means of energytransmission and reception. The antenna is preferably of a small size toallow it to fit into a mobile, handheld device where the available spacefor the antenna may be limited. An embodiment describes a highefficiency antenna for the specific characteristics and environment forthe power being transmitted and received.

Antenna theory suggests that a highly efficient but small antenna willtypically have a narrow band of frequencies over which it will beefficient. Many of skill in the art have, therefore, avoided the use ofthese antennas, to enable more flexible transmit and/or receivecharacteristics. In an embodiment, an adaptive tuning circuit is used incertain configurations to allow tuning of an efficient yet narrowbandantenna.

One embodiment describes an efficient power transfer between twoantennas by storing energy in the near field of the transmittingantenna, rather than sending the energy into free space in the form of atravelling electromagnetic wave. This embodiment increases the qualityfactor (Q) of the antennas.

This can reduce radiation resistance (R_(r)) and loss resistance(R_(l)).

In one embodiment, two high-Q antennas are placed such that they reactsimilarly to a loosely coupled transformer, with one antenna inducingpower into the other. The antennas preferably have Qs that are greaterthan 1000.

Another embodiment describes maximum Permissible Exposure (MPE) wherethe maximum exposure limits are defined by European and US standards (aswell as others). They are defined in terms of power density limits(W/m²), magnetic field limits (A/m) and electric field limits (V/m). Thelimits are related through the impedance of free space, 377 W.

In the US, the applicable standard is FCC CFR Title 47: §2.1091Radiofrequency radiation exposure evaluation: mobile devices. A mobiledevice is a transmitting device designed to be used in such a way thatthe separation distance of at least 20 cm is normally maintained betweenthe transmitter's radiating structure(s) and the body of the user ornearby persons. The limits to be used for evaluation are specified in§1.310 of Title 47.—§1.1310 Radiofrequency radiation exposure limits(see Table 1).

Table 1: FCC limits for radiation exposure Limits FOR MAXIMUMPERMISSIBLE EXPOSURE (MPE)

TABLE 1 FCC limits for radiation exposure LIMITS FOR MAXIMUM PERMISSIBLEEXPOSURE (MPE) Electric Magnetic Frequency field field Power Averagingrange strength strength density time (MHz) (V/m) (A/m) (mW/cm²)(minutes) (A) Limits for Occupational/Controlled Exposures 0.3-3.0  6141.63 *(100)   6 3.0-30   1842/1 4.691 *(300.1) 6 30-300 61.4 0.163 1.0 6300-1500 1/300  6   1500-100,000 5   6 (B) Limits for GeneralPopulation/Uncontrolled Exposure 0.3-1.34 614 1.63 *(100) 30 1.34-30  824/1 2.191   *(180.f³) 30 30-300 27.5 0.073 0.2 30 300-1500 1/1500 30  1500-100,000 1.0 30 f = frequency in MHz * = Plane-wave equivalentpower density NOTE 1 TO TABLE 1: Occupational/controlled limits apply insituations in which persons are exposed as a consequence of theiremployment provided those persons are fully aware of the potential forexposure and can exercise control over their exposure. Limits foroccupational/controlled exposure also apply in situations when anindividual is transient through a location where occupational/controlledlimits apply provided he or she is made aware of the potential forexposure. NOTE 2 TO TABLE 1: General population/uncontrolled exposuresapply in situations in which the general public may be exposed, or inwhich persons that are exposed as a consequence of their employment maynot be fully aware of the potential for or can not exercise control overtheir exposure.

In Europe, the applicable standard is EN60215. This has been derivedfrom the ICNIRP (International Commission on Non-Ionizing RadiationProtection) guidelines [ICN], The limits are given in Table 2.

Table 2: European limits for radiation exposure

TABLE 2 European limits for radiation exposure Table 2: European limitsfor radiation exposure Electric Magnetic Frequency field field PowerAveraging range strength strength density time (MHz) (V/m) (A/m) (W/m²)(min) 0.15-1     87 0.73/f — 6 1-10  87/f^(1/2) 0.73/f — 6 10-400  280.073 2 6 400-2000 1375 f^(1/2) 0.0037 f^(1/2) f/200 6   2000-300,000 61 0.16 10 6

The power density limits and magnetic field limits are of particularinterest in one embodiment. Using the data from Table 1 and Table 2,limit curves can be determined. FIG. 1 shows a plot of power densitywith the FCC limit curve 100, and the EN curve 102. FIG. 2 shows a plotof maximum H field, with the FCC curve 200, and the EN curve 202.

FIGS. 1 and 2 illustrate how the US limits are more generous atfrequencies below 30 MHz and could offset the effect of reduced antennaefficiency at low frequency. This study tests a range of frequencies tosee which frequencies are the best for wireless power transfer.

This application also provides an exemplary theoretical analysis ofvarious aspects of wireless energy and power transfer.

Embodiments disclosed herein describe antenna types.

A loop antenna is a “magnetic” antenna and may be less sensitive tochanges in its surroundings than a dipole, which is an “electric”antenna. The loop antenna may have certain advantages when the device isexposed to changes in its surroundings, e.g., when placed on a table,held in the hand, or put in a pocket, based on stray capacitance, orother effects. In an embodiment, an air loop antenna is used. Anotherembodiment may use a loop antenna with a ferrite core, or others may beused.

In one embodiment, an air loop antenna may be preferred over a loopantenna with a ferrite core. The air loop antenna may be more resistantto detuning from permanent magnets or other magnetic influences in itsvicinity. The air loop antenna will, in general, be more efficient thanthe ferrite loop antenna, since the ferrite core can cause losses. Theferrite antenna is often heavier, and cannot typically have componentsplaced “inside” it. In contrast, other components can be placed insidethe loop of an air loop antenna. The form-factor of the loop may bemodified or otherwise adapted to fit within a form-factor of certainportable devices being charged.

The same type of antenna may be used for both transmitter and receiver.The transmit and receive antennas can be the same or different sizes.

An embodiment describes a tuning circuit that becomes part of theantenna circuit. A typical loop antenna is inherently inductive. Acapacitive element may be used in the tuning circuit to induce resonancein the antenna. Even though a loop antenna is less sensitive to changesin its surroundings than a dipole antenna, it will still be detuned tosome degree by changes in its surroundings. Therefore, it may bedesirable in certain embodiments to adaptively tune either thetransmitter antenna or the receiver antenna or both, to maintain thequality of the link therebetween.

Adaptive tuning is achieved in one embodiment by changing the value of acapacitive element used in series with the loop antenna to adjust theresonant frequency of the circuit. The adaptive tuning circuit at thetransmitter and/or at the receiver. A goal is to choose tuningcomponents with high quality factors (Q) to ensure that that the Q ofthe overall receiver circuit is degraded as little as possible. In anembodiment, high Q is used to maximize efficiency, even at the cost ofnarrow bandwidth.

FIG. 3 illustrates an air loop antenna of an embodiment. The antenna mayhave maximum dimensions of 5 cm (i.e. a radius r of 2.5 cm) and N turnsof wire 300 of diameter 2b_(rx)=500 um is used for one embodiment. Theantenna can, for example, be placed around the perimeter of a mobiledevice. The loop will be assumed circular, but can be of other shapes. Acapacitor 302 is used with the loop inductive resonator to bring theloop antenna to resonance. The capacitor value may be defined as:

$\omega^{2} = \frac{+ 1}{LC}$

By calculating the inductance of the air loop antenna using the Equationand a wire diameter of 500 um, the required capacitance can becalculated for any of a number of frequencies.

In one embodiment, the capacitor 302 can be a high Q fixed chipcapacitor in parallel with a high Q varactor diode operating as avoltage-tunable capacitor to bring the receiver air loop to resonanceand to maintain tuning. FIG. 4 shows a schematic of the series-resonantcircuit formed with tuning circuit 400, that itself is formed ofvariable capacitance 402, in series with its equivalent seriesresistance 404. A fixed capacitor is shows as 406 in series with its ESR408. The antenna overall ohmic resistance 410 is shown separated asradiation resistance 410 in series with overall antenna resistance 412.Load resistance 414 and inductance 416 are also shown.

In the circuit, the symbols have the following meanings:

-   -   V₀: The induced voltage across the loop antenna    -   L_(rx): The inductance of the loop antenna    -   R_(l) _(—) _(rx), R_(r) _(—) _(rx), R_(a) _(—) _(rx):        Receive-antenna loss (ohmic) resistance, radiation resistance        and overall antenna resistance (the sum of the previous two)    -   C_(var), R_(esr) _(—) _(var): The capacitance of the tuning        varactor and its associated Equivalent Series Resistance (ESR)    -   C_(fix), R_(esr) _(—) _(fix): The fixed capacitance and its        associated ESR    -   R_(load) _(—) _(rx): The load resistance

One embodiment selects a tuning range of roughly +/−5 percent of thechosen operating frequency, so as to cover variations in the capacitanceand detuning from external factors. The varactor's tuning range could beapproximately +/−10 percent the fixed capacitance value. Components usedpreferably have a high Q, so that they degrade the overall Q of thecircuit as little as possible.

In one embodiment, tuning is carried out solely at the transmitter. Inthis embodiment, no varactor diode need be located at the receiver andthe transmitter tracks the receiver resonant frequency. This isdependent on how much resonant frequency of the receiver loop isaffected by changes in the environment near the loop. The oppositeconfiguration can also be used.

At higher frequencies, or with larger loop dimensions, or with more loopturns, a very small capacitance may be required to bring the loop toresonance. In an embodiment, only a varactor diode or only a fixedcapacitor would be used without the other.

Another effect to be considered is the self-resonance of the loop,especially at higher frequencies. This effect will occur asinter-winding capacitance and stray capacitances on the loop antennacome into resonance with the inductance of the winding itself. Thisdecreases as frequency increases.

At a lower operating frequency such as 1.3 MHz, a larger fixed capacitorwill be required. For example, the loop antenna with the dimensionsgiven in FIG. 3 with 5 turns of loop antenna would require a fixedcapacitance of about 3 nF. Capacitance variations of +/−1 percent (30pF) are typical for these types of capacitors. As will be shown, thisexceeds the tuning range of most available tunable capacitors.Therefore, at low frequencies, one embodiment locates the adaptivetuning only in the transmitter.

Increasing the operating frequency or increasing the number of turnsallows reducing the size of the fixed capacitance. A larger number ofturns may make the packaging more difficult. Therefore, with a largenumber of turns, practical implementation for certain types ofapplications could become difficult. A higher frequency therefore mightallow certain benefits in applications where this factor might otherwisebe limiting.

However, at frequencies of 250 MHz and above, the size of fixedcapacitor required is extremely small—e.g. on the order of 1 pF for N=1,and even less for more turns. At these frequencies, the fixed capacitorcan be eliminated altogether in some cases, and only a very small tuningcapacitor used. This physical limit on capacitor size also places alimit on the frequencies that can be used, for given loop dimensions. Asmaller receiver loop size would allow a higher frequency or more loopsto be used.

Exemplary high Q/low ESR capacitors with capacitances from the lowpicofarad to the low nanofarad range can be obtained commercially, e.g,from AVX Corp. Details of some potentially suitable AVX capacitors aretabulated in Table 3, although any number of other devices may be used.

Table 3

TABLE 3 Capacitor Capacitance Family Range of Voltage (all AVX) FamilyTolerance Q ESR Rating Dimensions HQ series, 3.3 pF to +/−0.25 pF Variesaccording Varies according 600 V to 9.4 mm × E case 6800 pF to +/−1% tocapacitance to capacitance 7200 V 9.9 mm × and frequency - andfrequency - 3.3 mm see FIG. 19 see FIG. 20 SQ AQ or 0.1 pF to +/−1%Claimed greater Approx 0.004 at 50 V 2.79 mm × CDR¹ 5100 pF than 10000at 1 1 MHz 2.79 mm × series, style MHz 2.59 mm 13 or 14

Note that in general, Q ESR and C are related by the following equation:

$C = \frac{1}{\omega \; R_{esr}Q}$

Another embodiment uses MEMS (Microelectromechanical Systems) varactors.This may lower the power consumption.

The circuit of FIG. 4 at resonance is analyzed to evaluate performance.In a first approach, the varactor will be replaced by a fixed valuecapacitor one-tenth the size of the main fixed capacitor. The AVX datawill be used for both capacitors. The tuning circuit 100 in FIG. 4 ismodeled as a single R/C impedance. Values used are:

I_(rx) is the current in the receiver loop

P_(rx) is the power at the load resistor

C_(ser) is the equivalent series capacitance of the fixed capacitor andvaractor, and

R_(ser) is the equivalent series resistance of the fixed capacitor andvaractor.

At resonance, the reactances can be neglected since XL=−Xc. Only theresistive (real) losses in the circuit are considered.

The inventors found that when the resistances of the tuned antenna arematched to the load resistance, the maximum amount of power P_(rx) isavailable at the load. In other words, the optimum condition is when

RL_(—rX)+Rr_(—rx)+R_(ser)=Rload_(—rx). The values may vary by 20% whilestill staying within “optimum” resonance.

In the embodiment, therefore, the transmitter circuit is modeled as aresonant loop where the loop is power-matched to the source. Anexemplary air loop antenna with maximum dimensions of 20 cm (i.e. aradius r of 10 cm), a wire radius of 1 mm and a single turn (N=1) isused for an embodiment, although other types, sizes and dimensions ofantenna may be used for the transmitter in other embodiments.

In one embodiment, the transmitter antenna could, for example, sitvertically on a bench or a table inside a home, within a wall, around awall power outlet, on or within a garage floor, behind a refrigerator,etc. To simplify calculations, the loop will be assumed circular, asFIG. 3. A single wire loop transmitting antenna of FIG. 3, having a wirediameter of 10 cm radius, wire radius of 1 mm, has an inductance ofapproximately 840 nH. Different frequencies will require differentcapacitance values for resonance with this antenna. For example, 1.3 Mhzwill require a capacitor of 17.85 nF; 13.56 Mhz will require 164.1 pF;64 Mhz will require 7.365 pF; 250 Mhz will require 0.483 pF and 500 Mhzwill require 0.121 pF.

A number of different antennas are described as embodiments herein. Fortesting of the embodiments, the antennas were built of 1.5 mm² copperwire and fixed onto a wooden frame.

The transmit antenna has a radius of 0.2 m, 6 turns, and 3 Mhz operatingfrequency. The matching is realized with two tunable capacitors. Thereceiving antenna has a radius of 0.1 m. Before considering powertransfer/pathgain, the antennas were tuned and measured independently.The resulting characteristics are summarized in Table 6.

TABLE 6 Transmitting Receiving antenna (TX) antenna (RX) Measuredunmatched characteristics @ 3 MHz R [Ω] 23 1.7 L [μH] 43 12.5 Usedmatching network C_(s) [pF] 36.5 33 C_(p) [pF] 27.6 187 Match [dB] −24−24 3 dB-bandwidth [kHz] 34 90 Quality factor Q 89 33

A quality factor increase may further increase power transfer. Thequality factor, for example, can be increased using a Matlab simulation.The mathematical investigations done for the simulation, leads to thefollowing approximation for the path gain:

${\mspace{14mu} {\eta (x)}} = \frac{\pi^{2}a^{6}Q_{ul}^{2}}{16{x^{6}\left\lbrack {\ln \left( {{8\frac{a}{b}} - 2} \right)} \right\rbrack}^{2}}$

-   -   where:    -   a=loop radius [m]    -   b=wire radius [m]    -   Q_(ul)=unloaded quality factor    -   x=distance between transmitter and receiver antenna [m]

The Equation above shows that for a practical antenna, the loop radiushas a high impact on the path gain.

SECOND EMBODIMENT ANTENNAS

FIG. 5 illustrates a second embodiment of the antennas. This embodimentobtains a maximized power transfer between a coupling loop 500 to whichthe transmitted power 502 is delivered. The coupling loop radiates to aresonator 510 with optimized Q. This embodiment uses a coupling loopwhich acts as a resonator instead of a coupling network made of twocapacitors. This reduces losses by omitting a matching network. Thecoupling between the coupling loop and the antenna can be conceptualizedas an ideal transformer.

The antenna may be made out of e.g., copper tube or the like in order todecrease the loss resistance by increasing the wire surface. Inaddition, the surface may be plated with silver (Ag) or another suchhigh conductance material well known in the art. With this type ofconstruction, a quality factor in the order of 1 is achieved, asdescribed in greater detail below. The resonator part of the antenna mayalso be optimized for a high quality factor (Q). This is done byincreasing the number of turns, increasing the surface of the wire andreducing dielectric losses due to isolation or the mounting of theantenna.

To tune the resonance frequency of the antennas, tunable capacitors 504,512 may be integrated at the bottom of both antennas. The capacitors maybe metal plates that are tunable by using three screws 514 to change thedistance between the two plates of the capacitor. The capacitorsdominate the self-capacitance (CS) of the antennas. Table 6A illustratesthe characteristics of these antennas.

TABLE 6A Transmitting Receiving antenna (TX) antenna (RX) Radius [m]0.085 0.085 Length [m] 0.078 0.078 Number of turns 7 7 Operatingfrequency [MHz] Resonance Resonance

The exemplary embodiments of the antennas are built of copper tube withan outer diameter of 6.0 mm. The surface is silver-plated. This protectsthe copper from corrosion and slightly increases the conductivity of thesurface.

With an exemplary plate-distance of the tuneable capacitor of 8 mm, theresulting calculated resonance frequency is 14.4 MHz.

Using a Q of 1300, pathgain is approx. −10 dB, at 1 m, which correspondsto a factor of 0.1. In other words, a transmitting power of 10 wattsmust be used to receive 1 W at the receiver.

The system should be defined around the unloaded Q (Q_(ul)) of theantennas. starting with:

$\begin{matrix}{{Q_{ul} = {\frac{1}{R} \cdot \sqrt{\frac{L}{C}}}},} & {{Equation}\mspace{14mu} 1\text{-}1}\end{matrix}$

The total loss resistance of either the Tx or Rx antenna can be definedby:

$\begin{matrix}{R = {\frac{1}{Q_{ul}} \cdot {\sqrt{\frac{L}{C}}.}}} & {{Equation}\mspace{14mu} 1\text{-}2}\end{matrix}$

At resonance, it can be written as

P _(in) =I ² ·R.  Equation 1-3

The resulting current in the TX-antenna can now be specified by

$\begin{matrix}{I = {\sqrt{\frac{P_{i\; n}}{R}}.}} & {{Equation}\mspace{14mu} 1\text{-}4}\end{matrix}$

Using Equation 1-2, the current can be rewritten as

$\begin{matrix}{I = {\sqrt{P_{i\; n} \cdot Q_{u\; l} \cdot \sqrt{\frac{C}{L}}}.}} & {{Equation}\mspace{14mu} 1\text{-}5}\end{matrix}$

The magnitude of the H-field generated by the current in the TX-antennain a distance x is

$\begin{matrix}{{{H(x)} = \frac{r_{A}^{2} \cdot I \cdot N}{2 \cdot \sqrt{\left( {r_{A}^{2} + x^{2}} \right)^{3}}}},} & {{Equation}\mspace{14mu} 1\text{-}6}\end{matrix}$

and induces a voltage

U _(ind)(x)=2πƒ_(res) ·N·πr _(A) ²·μ₀ ·H(x)  Equation 1-7

-   -   in the RX-antenna. The parameter r_(A) is the radius, N the        number of turns of the loop-antenna. The available output power        P_(out) can now be calculated with

$\begin{matrix}{{P_{out}(x)} = {\frac{{U_{ind}(x)}^{2}}{4 \cdot R}.}} & {{Equation}\mspace{14mu} 1\text{-}8} \\{{P_{out}(x)} = {{U_{ind}(x)}^{2} \cdot \frac{Q_{ul}}{4} \cdot {\sqrt{\frac{C}{L}}.}}} & {{Equation}\mspace{14mu} 1\text{-}9}\end{matrix}$

Finally, pathgain is defined as

$\begin{matrix}{{\eta (x)}_{dB} = {10 \cdot {{\log_{10}\left( \frac{P_{out}(x)}{P_{i\; n}} \right)}.}}} & {{Equation}\mspace{14mu} 1\text{-}10}\end{matrix}$

-   -   To further simplify and understand the behaviour of Equation        1-10 and Equation 1-9, models for L and C are needed. The        capacitance can simply be defined over the resonance frequency

$\begin{matrix}{C = {\frac{1}{\omega_{0}^{2} \cdot L}.}} & {{Equation}\mspace{14mu} 1\text{-}11}\end{matrix}$

-   -   For the inductivity, an empiric formula was found to be the most        accurate for the type of antenna used in this system.

$\begin{matrix}{L = \frac{\mu \; {\pi \cdot N^{2}}r_{A}^{2}}{{0.9\; r_{A}} + l_{A}}} & {{Equation}\mspace{14mu} 1\text{-}12}\end{matrix}$

The parameter I_(A) is the width of the antenna.

Under the assumption that the separation x between the antennas is largecompared to the radius of the antennas rA (x>rA), and with Equation 1-11and Equation 1-12, Equation 1-10 can be written as:

$\begin{matrix}{{\eta (x)}_{dB} = {10 \cdot {{\log_{10}\left( \frac{r_{A}^{2} \cdot {Q_{ul}^{2}\left( {{0.9\; r_{A}} + l_{A}} \right)}^{2}}{16\; x^{6}} \right)}.}}} & {{Equation}\mspace{14mu} 1\text{-}13}\end{matrix}$

The term in brackets in 1-13 is the linear pathgain. Note that thislinear pathgain is not a direct function of the frequency or the numberof turns, although these parameters are implicitly contained in thequality factor. The pathgain is approximately proportional to loopradius rA⁶, if the loop radius is much larger than the loop lengthl_(A). It is inversely proportional to the separation x⁶ between theantennas. It is also proportional to the quality factor Q_(ul) ².

For a given antenna dimension, as the quality factor is increased, thepathgain is improved. This is validated in an embodiment via simulation.The above equations were simulated using Matlab® to test antennas withdifferent sizes and quality factors. The following parameter set wasdefined to run the script:

% Parameter definitions

Q=1000; % target unloaded quality factor [1]

N=7;% number of turns [1]

r_loop=85e-3; % radius of loop antenna [m]

r_wire=3e-3; % radius of wire [m]

pitch=12e-3; % distance between two turns (center to center) [m]

freq=13.0e6; % system frequency [Hz]

dist=1:0.1:3; % distance of antennas [m]

P_in=1% input power [W]

The resulting simulation showed a pathgain variation

of −60 dB per decade which is caused by the term x⁶ in Equation 1-13. IfQ is doubled, for example from 1000 to 2000, pathgain increases by 6 dB.If the distance is doubled, pathgain decreases by 18 dB. The exemplarydefined parameters are valid for both TX- and RX-antennas, and hence canassist with forming an optimal antenna for the parameters.

The simulation also calculates the reactive voltages. The reactivevoltages occurring at the inductance and the capacitance are directlyproportional to the quality factor and proportional to the square rootof the transmitting power as set forth in Equation 1-14.

U _(LC) =Q _(ul)√{square root over (P _(in) ·R)}  Equation 1-14

Both reactive voltages will be very high in a practical implementation,thus planning for those voltages becomes more critical. With a Q of 1000and a transmitting power of 10 W, the voltages may be 2.7 kV. If a platecapacitor is used with a plate distance of 0.01 m, the resulting fieldstrength is 270 kV/m. Capacitors that can withstand these high voltagesbecome critical. For example, it may be necessary to use 2000 v or 3000V or higher voltage withstanding capacitors. It is believed that atleast one reason why systems of this type did not operate properly inthe past is that they were not properly sized for the amount of reactivevoltage at was actually present. In fact, the unexpectedly high voltageabove 2 KV is found as part of these reactive voltages even when muchsmaller voltages are being transmitted. This unexpectedly high voltageneeds to be handled by the circuit components.

Definition of the Quality Factor also becomes important, because a majorfocus of the antenna design process is on optimizing the quality factor.Accordingly, the following describes an in-depth analysis of Q.

The fundamental equation about the quality factor is given by Equation1-1 above.

$\begin{matrix}{{Q_{ul}} = {\frac{1}{R} \cdot {\sqrt{\frac{L}{C}}.}}} & {{Equation}\mspace{14mu} 1\text{-}1}\end{matrix}$

FIG. 6 illustrates a plot of Q vs frequency for a number of differentfrequencies.

Note that the proportion of L and C is important in this equation. For agiven resonance frequency, there are an infinite number of possible L-Ccombinations. However, higher Q is obtained when the inductance L is ashigh as possible compared to the capacitance.

The quality factor is also inversely proportional to the resistance R.This resistance consists of a loss resistance (R_(l)) and a radiationresistance (R_(r)). Both should be minimized in order to increase thequality factor.

The loss resistance is dependent of the material used to build theantenna, and due to the skin effect of frequency used for the system. Ahighly conductive material with good skin effect is preferable.

A high resonance frequency increases losses and hence decreases thequality factor. This is why the curve of FIG. 6 decreases at the upperend of the frequency-scale. However, a lower resonance frequency isobtained by increasing the capacitance. This decreases the L/C ratio,and since L is independent of the frequency, this lowers Q. Hence, theFIG. 6 curve shows how Q decreases at both the upper and lower end ofthe frequency-scale, making an ideal quality factor around a frequencyof 29 MHz for the given antenna dimensions.

This shows an ideal frequency or frequency range for each antennageometry.

The resonance frequency of 13 MHz used during testing described hereinis below this ideal frequency. This is because self resonance, which isthe resonance frequency without the tunable capacitor, below theresonator of the antennas is around 35 MHz. If the resonator is used atthis frequency without the tunable capacitor, the sensitivity of theantenna against close-by objects may become significant.

An embodiment minimizes this effect and at the same time makes itpossible to change the resonance frequency. A tunable capacitor of avalue which dominates the self-capacitance of the resonator is used forthis purpose. The added capacitance lowers the resonance frequency ofthe antenna.

Quality factor typically cannot be measured directly. Instead, thedefinition

$\begin{matrix}{Q = \frac{\omega_{0}}{\Delta \; \omega}} & {{Equation}\mspace{14mu} 2\text{-}1}\end{matrix}$

may be used as a starting point, where ω₀ is the center or resonancefrequency and Δω corresponds to the 3 dB-bandwidth. Q can therefore befound by measuring the two parameters ω₀ and Δω.

The 3 dB-bandwidth can be found as follows. The impedance Z of a firstorder series RLC-circuit is given by

$\begin{matrix}{{\mspace{14mu} \underset{\_}{Z}} = {R + {j\; \omega \; L} + {\frac{1}{j\; \omega \; C}.}}} & {{Equation}\mspace{14mu} 2\text{-}2}\end{matrix}$

-   -   With the help of

$\begin{matrix}{\omega_{0} = \frac{1}{\sqrt{LC}}} & {{Equation}\mspace{14mu} 2\text{-}3} \\{and} & \; \\{{Q = {\frac{1}{R} \cdot \sqrt{\frac{L}{C}}}},} & {{Equation}\mspace{14mu} 2\text{-}4}\end{matrix}$

-   -   the inductance L and capacitance C can be written in terms of Q        and ω₀.

$\begin{matrix}{L = \frac{QR}{\omega_{0}}} & {{Equation}\mspace{14mu} 2\text{-}5} \\{C = \frac{1}{{QR} \cdot \omega_{0}}} & {{Equation}\mspace{14mu} 2\text{-}6}\end{matrix}$

-   -   If Equation 2-5 and Equation 2-6 is used in Equation 2-2,        impedance can be expressed with

$\begin{matrix}{\underset{\_}{Z} = {R + {{j \cdot {QR}}{\left( {\frac{\omega}{\omega_{0}} - \frac{\omega_{0}}{\omega}} \right).}}}} & {{Equation}\mspace{14mu} 2\text{-}7}\end{matrix}$

-   -   The quality factor Q can also be used to define the bandwidth        (like in Equation 2-1)

$\begin{matrix}{\frac{\Delta \; \omega}{2} = {\frac{\omega_{0}}{2Q}.}} & {{Equation}\mspace{20mu} 2\text{-}8}\end{matrix}$

The impedance phase is given by the inverse tangent of the imaginarypart of Z, divided by the real part of Z. In this division, R cancelsout. If in addition Equation 2-8 is used and the function is evaluatedat the upper cut-off frequency, then the phase is given by

$\begin{matrix}{{\phi \left( {\omega_{0} + \frac{\Delta \; \omega}{2}} \right)} = {{\tan^{- 1}\left( {Q\left( {\frac{\omega_{0} + \frac{\omega_{0}}{2Q}}{\omega_{0}} - \frac{\omega_{0}}{\omega_{0} + \frac{\omega_{0}}{2Q}}} \right)} \right)}.}} & {{Equation}\mspace{20mu} 2\text{-}9}\end{matrix}$

-   -   If the expression in the bracket is simplified, the phase gets        dependent only from Q.

$\begin{matrix}{{\phi \left( {\omega_{0} + \frac{\Delta \; \omega}{2}} \right)} = {{\tan^{- 1}\left( {Q + \frac{1}{2} - \frac{Q}{1 + \frac{1}{2Q}}} \right)}.}} & {{Equation}\mspace{20mu} 2\text{-}10}\end{matrix}$

If Q increases, the function in the bracket tends 1.

$\begin{matrix}{{\phi \left( {\omega_{0} + \frac{\Delta \; \omega}{2}} \right)} = {{\tan^{- 1}(1)} = {\frac{\pi}{4}.}}} & {{Equation}\mspace{20mu} 2\text{-}11}\end{matrix}$

The result from Equation 2-11 corresponds to an angle of 90 degreeswhich implies that the imaginary part of Z is equal to the real part ofZ. Those two points can then be found with a network analyzer, and thenEquation 2-1 can be used to calculate Q. Hence, using this framework,the Q value of such an antenna can be actually determined.

A second embodiment of the antenna is shown in FIG. 7. A coupling loop700 is placed approximately 0.1 m away from the main part of the antenna710. A plate capacitor is formed between two copper plates 721, 722.Screws 723, 724, 725 are formed of a capacitively-inert material such aspolyimide. These screws are used to adjust the capacitance provided bythe tuneable capacitor 720 and in turn adjust the resonance frequency ofthe antenna.

A glass body 730 or other dielectric may be below the antenna tominimize losses due to obstacles below the antenna.

Moreover, as described above, surface conduction is important tomaximize Q. A silver plating or other non-corrosive material, may beapplied to protect the copper from corrosion.

The coupling loop 700 may be formed of the same copper tube material,but has only one turn and about half the diameter of the antenna. Thecoupling loop is placed around 0.1 m away from the antenna to get a 50 Wmatching.

The following explains how the resonance frequency of the antenna partscan be determined. In the following equations, L is the inductance ofthe resonator itself, Cs is the self-capacitance of the resonator and CTis the tuneable capacitor associated with the resonator, Rr is theradiation resistance, Rl the loss resistance of the resonator.

$\begin{matrix}{{L = \frac{\mu_{0}{\pi \cdot N^{2}}r_{A}^{2}}{{0.9r_{A}} + l_{A}}}{{where}\text{:}}{r_{A} = {{loop}\mspace{14mu} {radius}}}{N = {{number}\mspace{14mu} {of}\mspace{14mu} {turns}}}{l_{A} = {{length}\mspace{14mu} {of}\mspace{14mu} {antenna}}}{\mu_{0} = {1.2566 \cdot 10^{- 6}}}} & {{Equation}\mspace{20mu} 3\text{-}1} \\{{{C_{S} = \frac{{\pi^{2} \cdot 2}{r_{A} \cdot ɛ_{0}}}{\ln\left( {\frac{p}{2r_{A}} + \sqrt{\left( \frac{p}{2r_{A}} \right)^{2} - 1}} \right)}}{{where}\text{:}}{{p = {{pitch}\mspace{14mu} {of}\mspace{14mu} {the}\mspace{14mu} {antenna}}},\mspace{45mu} {{corresponds}\mspace{14mu} {to}\mspace{14mu} {the}\mspace{14mu} {distance}\mspace{14mu} {between}}}\mspace{14mu} \mspace{45mu} {{two}\mspace{14mu} {turns}\mspace{14mu} {plus}\mspace{14mu} {the}\mspace{14mu} {diameter}\mspace{14mu} {of}\mspace{14mu} a\mspace{14mu} {turn}}{ɛ_{0} = {8.8542 \cdot 10^{- 12}}}}{{Note}\text{:}\mspace{20mu} {{see}\lbrack{GRA}\rbrack}\mspace{14mu} {for}\mspace{14mu} a\mspace{14mu} {derivation}\mspace{14mu} {of}\mspace{14mu} {this}\mspace{14mu} {formula}}} & {{Equation}\mspace{20mu} 3\text{-}2} \\{{C_{T} = \frac{ɛ_{0} \cdot A}{d}}{{where}\text{:}}{A = {{area}\mspace{14mu} {of}\mspace{14mu} {the}\mspace{14mu} {plate}\mspace{14mu} {capacitor}}}{d = {{distance}\mspace{14mu} {between}\mspace{14mu} {the}\mspace{14mu} {plates}}}{{Note}\text{:}\mspace{14mu} {this}\mspace{14mu} {is}\mspace{14mu} {an}\mspace{14mu} {approximate}\mspace{14mu} {formula}\mspace{14mu} {because}}\mspace{70mu} {{fringing}\mspace{14mu} {is}\mspace{14mu} {{neglected}.\mspace{11mu} {The}}\mspace{14mu} {real}\mspace{14mu} {capacitance}}\mspace{70mu} {{is}\mspace{14mu} {higher}\mspace{14mu} {than}\mspace{20mu} {the}\mspace{14mu} {{calculated}.}}} & {{Equation}\mspace{20mu} 3\text{-}3} \\{f_{0} = \frac{1}{2{\pi \cdot \sqrt{L \cdot \left( {C_{T} + C_{S}} \right)}}}} & {{Equation}\mspace{20mu} 3\text{-}4} \\{{R_{r} = {320\; \pi^{4}{N^{2}\left( \frac{\pi \cdot r_{A}^{2}}{\lambda^{2}} \right)}}}{{{where}\text{:}}\lambda = {{wavelength}\mspace{14mu} {at}\mspace{14mu} {the}\mspace{14mu} {frequency}\mspace{14mu} f_{0}}}} & {{Equation}\mspace{20mu} 3\text{-}5} \\{{R_{t} = {1.25 \cdot \frac{N \cdot \pi \cdot r_{A}}{r_{w}} \cdot \sqrt{\frac{f_{0} \cdot \mu_{0}}{\sigma \cdot \pi}}}}{{where}\text{:}}{\sigma = {{conductivity}\mspace{14mu} {of}\mspace{14mu} {the}\mspace{14mu} {metal}\mspace{14mu} {used}\mspace{14mu} {for}\mspace{14mu} {the}}}\mspace{50mu} {{antenna},{{45 \cdot 10^{6}}\mspace{14mu} S\text{/}m\mspace{14mu} {used}\mspace{14mu} {for}\mspace{14mu} {calculations}}}{r_{w} = {{radius}\mspace{14mu} {of}\mspace{14mu} {the}\mspace{14mu} {copper}\mspace{14mu} {tube}}}} & {{Equation}\mspace{20mu} 3\text{-}6}\end{matrix}$

Table 8 shows the values obtained for the current values of antennaparameters.

TABLE 8 L [μH] (calculated value) 9.05 C_(S) [pF] (calculated value) 1.9C_(T) [pF] (calculated value with the plate- 6.2-18.6 distance varyingfrom 5 mm-15 mm) R_(r) [Ohm] 0.0028 R_(l) [Ohm] 0.264 f_(R65) [MHz](measured) 12.5 Quality factor Q (calculated) 2780 Quality factor Q(measured) 1300

An exemplary test arrangement for the pathgain-measurement may becarried out to obtain the actual values. This measurement may completelydecouple the transmitter from the receiver, to avoid power transfer onthe surface of the coaxial shields and on the power lines. A signalgenerator and a battery powered spectrum analyzer may be used on thetransmitter and the receiver side respectively.

To measure the quality of the matching, the energy returning from thetransmitting antenna is measured with a power meter which was connectedthrough a directional coupler. The forward coupling port of thedirectional coupler was terminated with a 50 W load. During themeasurements, the matching was at least 20 dB. Matching can be varied byadjusting the distance between the antenna and the coupling loop.

On the receiver side, the spectrum analyzer is directly connected to thereceiving antenna.

The same resonance frequencies are used for both antennas. Detuningresults in a dramatically reduced power transfer. The receiving antennamay use a tuning aid, e.g., a Teflon bar that is selectively insertableinto the tunable capacitor of the antenna, resulting in aresonance-shift of approximately 40 kHz, or may use an adjustablecapacitor as previously described. For each distance measured, thereceiving antenna is tuned to receive the maximum power available. Thetransmitting antenna is tuned by slightly adjusting the signalgenerator's frequency.

Table 9 shows the resulting measured pathgains.

TABLE 9 Level at Level at Pathgain Distance [m] receiver [dBm]transmitter [dBm] [dB] 1.1 3.3 12.5 −9.2 1.5 −3.7 12.5 −16.2 2.0 −10.512.5 −23.0 2.5 −18.5 12.5 −31.0 3.0 −25.3 12.5 −37.8

Moreover, since the reactive voltage can easily exceed several kV, itmay be useful to test the antenna prototypes to determine that reactivevoltage, in order to allow determination of proper sizing for thecapacitors. An electrically decoupled system may be used. A sourcesignal from the signal generator is amplified by a 50 dB RF amplifier.The 20 dB attenuator in-between is used to limit the available power onthe TX antenna.

The 3 dB attenuator after the amplifier is used to protect the amplifierin case of a mismatched antenna. To measure the quality of the matching,a power meter is used to show the energy returning from the antenna. Onthe receiver side, a small light bulb (50 W/3 W) may be used to indicatethe received power. Tuning and matching was realized by using the tuningaid, by varying the frequency of the signal generator and by varying thedistance between the antennas and the coupling loop.

The results are summarized in the Table 10:

TABLE 10 Distance [m] 1.2 Pathgain in dB −11 Transmitting power [W] 10Receiving power [W] 0.8 Approximate reactive voltage [kV] 3.1Approximate field strength in the 310 tuneable capacitor [kV/m]

In one embodiment, sensitivity to close-by objects or humans may cause ashift of the resonance frequency due to e.g., stray capacitance. Thismay be mitigated by a shield, e.g., a slotted or other shield, disposedaround the antenna.

In another embodiment, a piece of mica which has a high dielectricstrength and very good isolating capabilities is used in place of thepolyimide screws noted above. This also raises the quality factor. It ispostulated suggesting that less energy is absorbed by the mica comparedto the polymer.

In another configuration, very thin pieces of mica or Teflon are used tohold the capacitor and limit transmitting power.

There is a tradeoff between Q and bandwidth. Due to the high qualityfactor, the bandwidth of the exemplary second antennas is somewhatnarrow, e.g., around 9 kHz at a resonance frequency of 13 MHz. Thisresults in certain tuning requirements, because both antennas have towork on almost exactly the same resonance frequency. Hence, in anotherembodiment, the antenna sensitivity to approaching objects as describedabove is reduced using the shield described above. An electronic tuningcircuit may be used to automatically tune the antenna circuit(s) tomaintain coherence. The detuning issue becomes especially important in asystem like this where the Q is very narrow. This narrow Q impliesnarrow bandwidth, which requires that the tuning be closer.

To further evaluate the effects of various external or design factors onQ, various aspects of the experimental setup are considered, includinge.g. the glass body under the antenna, the piece of mica in thecapacitor, the losses of the coupling loop, and the losses of all theobjects within the near field of the antenna.

To evaluate these factors, the environment of the antenna should be asideal as possible. Therefore, the exemplary antenna was suspended withtwo thin nylon strings. Two coupling loops are used to measuretransmission (S 12/S21) instead of reflection (S 11). The coupling loopswere placed on both sides of the antenna, about 0.6 m away from theantenna, to achieve an undercritical coupling. That is, when an equalamount of power is dissipated in the external circuit as in theresonator itself, the coupling is said to be critical (and the antennais matched). An undercritical coupling means that more power isdissipated in the resonator than in the external circuit, while anovercritical coupling means that more power is lost in the externalcircuit than in the resonator.

The theoretically expected Q for this embodiment is 2877. The realized Qof 2263 is 78.6% of the theoretical value. The theoretical value wasalmost reached in this test. It was also noted that higher Q, however,may make the antenna more susceptible to influence by its environment.Thus, in practice, the Q will likely always be lower than thetheoretical value.

A second measurement showed another characteristic of the qualityfactor. The loaded quality factor QL should be half of the unloadedquality factor (Qul) under the condition when the resonator iscritically coupled.

The foregoing transmitter and receiver apparatus (e.g., antenna. and anyassociated electronic or electrical components) may also be combined inanother embodiment as a transceiver: i.e. a device adapted to bothtransmit and receive power. This may comprise for example a common formfactor (e.g., a single unit or “box” having both transmit and receiveantennas and circuitry disposed therein). Moreover, the device may beused as a repeater to receive energy from one source via the receiveantenna, and then transmit the received power to another source (or backto the same on) using the transmit antenna. These events (transmissionand reception) may occur at the same time or with one delayed relativeto each other. Values can be modified depending on the polarization,strength, geometry, relative spacing and placement, and other factorsassociated with the transmit and receive antennas, or may also beconducted according to any number of well known multiple access schemessuch as e.g., frequency division or FDMA (e.g. wherein the resonantfrequency of the first antenna (receiver or transmitter) is different orseparated from that of the second antenna (transmitter or receiver). Asyet another option, the two antennas may use the same or differentfrequency, and be time-divided or slotted as to their operation (e.g.,TDMA).

In another alternative, a “CSMA” like approach may be used (whether withour without “collision detection”), such as where one device actively orpassively detects or senses the activity of the other, and adjusts itsbehavior accordingly. In one such embodiment, the transmitter, beforetransmitting, detects the state of the receiver (e.g., whether inresonance, generating current, etc.) and uses this as a gating criterionfor transmission).

Another embodiment uses a “resonant frequency hopping” approach, whereinmultiple access or other aims, such as defeating or mitigating Rayleighor antenna diversity fading or other such issues, is accomplished by wayof periodically or deterministically or psuedorandomly hopping theresonant frequency as a function of time. For example, the transmitterand receiver may “seed” corresponding deterministic algorithms so as tomutually generate a common hop sequence that allows them to maintainsynchronized. Alternatively. “in-band” (e.g., modulated power signal)signaling may be used to transmit the hop sequence in advance (or as itproceeds) to the receiver from the transmitter; e.g . . . . “I will hopto frequency X at next interval, and frequency X+Y after that . . . ”,and so forth. A separate low power transmitter, e.g., RF or Bluetooth,can be used to synchronize the specific information. Clockinginformation may also be sent in an analogous way.

In another embodiment, a passive “collision detection” or CD approach isused, such as where the transmitter attempts to transmit, and determineswhether an interfering operation is occurring at the same time. Forexample, the determination may be by detecting a resonant frequency, atransmission efficiency, a feedback from a receiver, or some otherdetection. This interfering operation may be caused by the operation ofthe receiver, a parasitic or stray capacitance effect, a loss of tuning,or other similar effect.

The transmitter may take an action at that point to avoid the issue.

In an embodiment, since the interference is typically temporary, thetransmitter can terminate the ongoing transmission and retry a latertime. One example of this is via a random backoff via an anti-collisionalgorithm. One embodiment allows the power has transmitted to be storedin a storage part such as a battery. Because of this, the device can thepower transmission can be stopped temporarily; while still allowing thepowered device to operate.

The transmitter can attempt to tune itself to a different resonantfrequency so as to mitigate the interference and/or attempt to tune orotherwise vary the operation of the receiver.

Another option is to increase the gain so as to increase an energytransfer rate. This may operate to “blast through” the interference asit were).

A system of adjusting polarization or orientation of transmitter and/orreceiver can be used, such as via a motor drive or similar mechanismthat physically alters the position of the antenna(s).

Any combination of the above can alternatively be used. These featurescan be implemented at the transmitter, and/or at the receiver basis, orin tandem or coordination between the transmitter and receiver.

Another embodiment uses signaling information between the devices thatrelates to the level or rate of power transfer as determined by thereceiver (e.g., “here's what I'm actually receiving, so you can comparethis to what you are actually sending to tune yourself, transmitter”).Moreover, the aforementioned multiple access schemes can be implementedin this fashion; e.g., backoff or CD based on a drop in receiverreceived power as communicated back to the transmitter via acommunication channel.

As yet another alternative embodiment, multiple transmitters and/orreceivers may be used, and the aforementioned features implemented asbetween the multiple transmitters (and/or receivers). For example, theFDMA, TDMA, CSMA or CD techniques may be applied as between twopotentially conflicting transmitters.

The foregoing functions may be implemented at an electrical orelectronic component level; e.g., via simple gate logic or the likeimplemented as anything from discrete components through highlyintegrated circuits, as computer programs or applications running one.g., a micro-controller or digital processor, via firmware disposed ona IC, manually, or in hardware to the degree applicable (e.g.electromechanical tuners, motors, etc.).

Moreover, the present application contemplates the dynamic alteration orvariation of one or more antenna or circuit parameters by environmentalcharacteristics. For example, this can change antenna characteristics.The characteristics that can be changed can include tuning capacitance,resistance value, radius of the loop or coil, e.g. via a thermal effectthat causes elongation or contraction of the material used to form theantenna loop(s), thereby changing its effective radius, the propertiesof the antenna or components in proximity thereto (e.g. by selectivelyapplying a particular electric field or magnetic field, alignment ofdipoles within the material or components might be selectively altered,thereby affecting the properties of the antenna), as well as electricalor electronic processing such as power signal processing, filtration,modulation, and others.

For example, in one embodiment, the (predominantly) magnetic field ofthe transmitter is modulated or impressed with information so as totransfer information, along with power.

In another embodiment this information comprises control signalingbetween the receiver and transmitter, thereby obviating any separatedata or communications link (and hence simplifying the system and makingit more robust). For instance, the transmitter may modulate thetransmitted narrowband signal as a function of time, e.g., via anamplitude modulation, phase modulation. sideband or frequencymodulation; e.g., GMSK or sideband shift up or down to encode a data “0”or “1”, pseudo-noise modulation. or other technique. This allowstransfer of information relating to inter alia transmitter parameters(such as resonant frequency. polarization, etc. that might otherwiseallow the receiver to better “lock on” to the transmitter to optimizepower transfer. Duty cycle, clock, or timing information might also beencoded: e.g., windows during which the transmitter will be operational,to synchronize time frames of reference etc.

The theoretical limits on antenna performance are explained herein.Aspects such as the antenna efficiency, quality factor (bandwidth) andsize are considered. In addition, a model for the radio wave propagationin the near-field and in the far-field is established.

An electrically small antenna is an antenna that can be fitted into afraction of a radiansphere, which is a sphere of radius rmax

$\begin{matrix}{r_{\max} = {\frac{1}{k} = {\frac{\lambda}{2\pi} = {\frac{c}{2\pi \; f} = \frac{d_{\max}}{2}}}}} & \left( {{eq}\mspace{20mu} A\text{-}1} \right)\end{matrix}$

where:

k is the wavenumber in m−1

l is the wavelength in m

c is the speed of light 299792458 ms−1

f is the frequency in Hz, and

dmax is the diameter of the radiansphere

The antennas used for this application will be electrically small inalmost all cases (i.e. kr<1), where kr=d/lo.

Electrically small antennas are not typically self-resonant. For lowfrequencies, the antennas are either capacitive (dipole antenna) orinductive (loop antenna). They can be approximated for example by afirst-order series RC or parallel RL circuit. They are brought toresonance by tuning with a reactor of an opposite kind.

The equivalent circuit of such an antenna is shown in FIG. 8 for thecapacitive case. One main element of the antenna is the radiationresistance Rr. In the equivalent circuit, this resistor models theradiated power. The loss resistor RL accounts for the conduction anddielectric losses of the antenna. The capacitor C represents thereactive component of the antenna. This, together with the externalmatching inductor L forms a resonant circuit, which is tuned to theoperating frequency. This circuit can also be modeled as an equivalentrepresentation as a parallel resonant circuit.

$\begin{matrix}{{{R_{o} + {{j\omega}_{o}L}} = {\left( \underset{\underset{R_{a}}{}}{R_{L} + R_{r}} \right) - {j\; \frac{1}{\omega_{o}C}}}},{\omega_{o} = {\frac{1}{\sqrt{LC}}{where}\text{:}R_{o}\mspace{14mu} {is}\mspace{14mu} {the}\mspace{14mu} {source}\mspace{14mu} {resistence}\mspace{14mu} {in}\mspace{14mu} \Omega R_{a}\mspace{14mu} {is}\mspace{14mu} {the}\mspace{14mu} {antenna}\mspace{14mu} {resistence}\mspace{14mu} {in}\mspace{14mu} \Omega R_{L}\mspace{14mu} {is}\mspace{14mu} {the}\mspace{14mu} {loss}\mspace{14mu} {resistence}\mspace{14mu} {in}\mspace{14mu} \Omega R_{r}\mspace{14mu} {is}\mspace{14mu} {the}\mspace{14mu} {radiation}\mspace{14mu} {resistence}\mspace{14mu} {in}\mspace{14mu} \Omega \omega_{o}\mspace{11mu} {is}\mspace{14mu} {the}\mspace{14mu} {resonance}\mspace{14mu} {frequency}\mspace{14mu} {in}\mspace{14mu} {rads}^{- 1}L\mspace{14mu} {is}\mspace{14mu} {the}\mspace{14mu} {matching}\mspace{14mu} {inductance}\mspace{14mu} {in}\mspace{14mu} HC\mspace{14mu} {is}\mspace{14mu} {the}\mspace{14mu} {antenna}\mspace{14mu} {capacitance}\mspace{14mu} {in}\mspace{14mu} F}}} & \left( {{eq}\mspace{20mu} A\text{-}2} \right)\end{matrix}$

For maximum power transfer, the antenna and matching network impedanceis complex conjugate matched at resonance to the antenna impedance.

A similar circuit can be derived for the case of an inductive antenna.

Applicants believe that there are fundamental limits on the efficiencyand quality Factor of such an antenna. If a certain antenna performanceis required, the size of an antenna cannot be reduced to an arbitraryvalue. Like the well-known Shannon limit in communication theory, whichrelates channel capacity to bandwidth and dynamic range, there is also afundamental limit in antenna theory that relates minimum antenna size tothe radiation quality factor.

There have been many attempts to calculate the theoretical limits onantenna size. The fundamental work was done by Chu [CHU] and Harrington[HAR]. Their theory states that the antenna is completely enclosed by asphere of radius r, The field outside the sphere, can be expressed as asum of weighted spherical waves propagating radially outward. Each wave(mode) exhibits power orthogonality and therefore carries powerindependently from the others.

It can be mathematically proven that a particular field outside thesphere can be generated with an infinite number of different sourcedistributions. The field outside the sphere is therefore independentfrom a particular implementation of the antenna. From Chu's calculationit has been shown that an antenna that excites only one mode (eitherTE01 or TM01) achieves the lowest possible radiation quality factor ofany linearly polarized antenna. Based on the above-described fundamentalwork. Hansen derived an approximate analytical expression for thisquality factor Qr [HAN], which has been cited many times in theliterature. Mclean further developed and corrected the work from Hansen[MLE], giving an exact expression for the radiation quality factor Qr ofa linear polarized antenna:

$\begin{matrix}\begin{matrix}{= \begin{Bmatrix}{{2\omega \; \frac{W_{e}}{P_{r}}},{W_{e} > {W_{m}\left( {{capactive}\mspace{14mu} {antenna}} \right)}}} \\{{2\omega \; \frac{W_{e}}{P_{r}}},{W_{m} > {W_{e}\left( {{inductive}\mspace{14mu} {antenna}} \right)}}}\end{Bmatrix}} \\{= {\frac{1}{({kr})^{3}} + \frac{1}{kr}}}\end{matrix} & {{Equation}\mspace{20mu} 3}\end{matrix}$

-   -   where    -   Q_(r) is the radiation quality factor (unitless)    -   ω is the radian frequency in rads⁻¹    -   W_(e) is the time-averaged, non-propagating stored electric        energy in J    -   W_(m) is the time-averaged non-propagating, stored magnetic        energy in J    -   P_(r) is the radiated power in W

This equation shows that the dominant term for electrically smallantennas (k; <<l) is the cubic term. However, for large antennas(k; >>l) the radiation quality factor will be governed by the linearterm.

A physical implementation of an antenna exhibits losses, i.e. itsradiation efficiency is smaller than unity due to non-ideal conductorsand dielectrics. The reduction of the efficiency has an impact on theoverall quality factor, called the antenna quality factor. Assuming theantenna is power-matched to the source, the antenna quality factor Qaresults in:

Q_(n)=η_(r)Q_(r)

-   -   where:    -   Q_(n) is the antenna quality factor (unitless)

where:

Q″ is the antenna quality factor (unitless)

Three important relations can be derived from Equation A3 and EquationA4:

-   -   For Small antennas the efficiency is proportional to the cube of        the relative antenna size and therefore also proportional to the        cube of the antenna size and to the cube of the frequency:

η_(r)∝(kr)²∝r³∝ƒ³  Equation 5

-   -   For large antennas the efficiency is proportional to the        relative antenna size and therefore also proportional to the        antenna size and the frequency:

η_(r)∝kr∝r∝ƒ  Equation 6

-   -   In general, the radiation efficiency is proportional to the        antenna quality factor:

η_(r)∝Q_(a)

  Equation 7

-   -   For the antenna models in FIG. 4 and FIG. 5 the values for        radiation quality factor Q_(r) and radiation efficiency η_(r)        are given as:

$\begin{matrix}{Q_{r,{cap}} = {\frac{{Im}\left\{ Z_{a} \right\}}{{Re}\left\{ Z_{a} \right\}} = \frac{1}{\omega_{o}R_{r}C}}} & {{Equation}\mspace{20mu} 8} \\{Q_{r,{ind}} = {\frac{{Im}\left\{ Y_{a} \right\}}{{Re}\left\{ Y_{a} \right\}} = \frac{R_{r}}{\omega_{o}L}}} & {{Equation}\mspace{20mu} 9} \\{\eta_{r} = {\frac{P_{r}}{P_{i\; n}} = \frac{R_{r}}{R_{r} + R_{L}}}} & {{Equation}\mspace{20mu} 10}\end{matrix}$

-   -   where:    -   η_(r) is the radiation efficiency (unitless)    -   Z_(n) is the antenna input impedance in Ω    -   Y_(a) is the antenna input admittance in Ω⁻¹    -   P_(r) is the radiated power at resonance in W    -   P_(in) is the power input to the antenna at resonance in W

This shows that for a given radiation efficiency, reducing antenna sizeleads to increased antenna quality factor. For a given antenna size,decreasing radiation efficiency results in lower antenna quality factor.Consequently, for a given radiation efficiency, a higher antenna qualityfactor is the penalty for a small antenna size.

The antenna quality factor decreases with increasing frequency andincreasing antenna size when the radiation for the wireless powering andcharging system the antenna efficiency is the most important criterion,as this determines how much power can be transmitted between twoantennas. Equation 5 illustrates that the antenna efficiency isproportional to the cube of the relative antenna size and therefore alsoproportional to the cube of the absolute antenna size. Increasing thesize by a factor of 10 results in an improvement of antenna efficiencyof 30 dB (factor 1000), assuming that the antenna quality factor is keptconstant.

Equation 7 shows that the antenna quality factor is proportional to theantenna efficiency. Increasing by 10 times the antenna quality factoryields an increase of the antenna efficiency of 10 dB (factor 10),assuming a constant relative antenna size. Antenna efficiency isproportional to the cube of the frequency. An increase by a factor of 10in the frequency leads to an improvement of the antenna efficiency by 30dB (factor 1000), assuming that the antenna size and the antenna qualityfactor are kept constant.

Unlike the fundamental limits on efficiency and quality factor that havebeen described above, the gain does not present a physical limit.However, as opposed to the gain, there is a good knowledge of thedirectivity that can be achieved with certain antenna types. Thedirectivity is linked to the gain as follows:

G=hD  (A11)

According to Balanis [BAL], the directivity of a small dipole is D=1.5.The same directivity applies also to a small loop. This similaritybecomes clear when the principle of duality of the electric and magneticfield is applied, as a small loop can be described with the help of amagnetic dipole.

Higher directivities can be expected from antennas that are notelectrically small. This is the case e.g. for the dipole as can be seenfrom FIG. A1. If the maximum antenna dimension is in the order of awavelength, the directivity is higher than that of the small dipole.However, for the wireless powering and charging system this is only thecase for frequencies above 1 GHz.

Radio Wave Propagation

The characteristics of an antenna show a strong dependence on the point(in terms of distance) where their fields are observed. A distinctionbetween near field and far field is often made. In the near-fieldregion, the electromagnetic energy is mainly stored and not radiated(stationary wave). The boundary for this region is usually defined as:

-   -   Near-field: In the near-field region the electromagnetic energy        is mainly stored and not radiated (stationary wave). The        boundary for this region is usually defined as:

$\left. {{kr}{\operatorname{<<}1}}\leftrightarrow{r{\operatorname{<<}\frac{\lambda}{2\pi}}} \right.,$

-   -   where;    -   k is the wave number, and    -   r the observation distance to the antenna.

In the far-field region most of the electromagnetic energy is radiatedand not stored. The boundary for this area is usually defined as:

-   -   Far-field: In the far-field region most of the electromagnetic        energy is radiated and not stored. The boundary for this area is        usually defined as:

${kr}\operatorname{>>}\left. 1\leftrightarrow r \right.\operatorname{>>}{\frac{\lambda}{2\pi}.}$

Between the near-field and the far-field a transition from a stationaryinto a propagating wave occurs. This is the so-called transition region.

For a distance of 0.5 to 5 m to the antenna the boundary between thenear-field and the far-field is in the frequency range of 10 to 100 MHz.

All radio waves propagate in a very different manner in the near-fieldand in the far-field. From radio communication theory the Friistransmission equation is well known. It describes the ratio of receivedpower to power of a transmit antenna. assuming a certain receive andtransmit antenna gain, as well as a certain separation between theseantennas:

$\begin{matrix}{\frac{P_{Rx}}{P_{Tx}} = {G_{Tx}{G_{Rx}\left( \frac{\lambda}{4\pi \; r} \right)}^{2}}} & {{Equation}\mspace{20mu} 12}\end{matrix}$

This equation is only valid in the Far-field. For a more generaltreatment of energy transmission between two antennas, a new equation isdeveloped that also covers the near-field.

The radiated fields of an electrically small dipole will be consideredAs a basis for this general radio wave propagation model. The dipole canalso be used to model a loop antenna because of the principle of dualityof the electric and magnetic field. Because of this, the electric fieldcomponent of a dipole corresponds to the magnetic field component of theloop and vice versa.

Equation 13 and Equation 14 show the components of the electric and themagnetic field of a small dipole. The radial component of the electricfield has been omitted, as it accounts only for the reactive energy thatis stored in the near-field.

$\begin{matrix}{E_{\theta} = {{j\eta}{{\frac{k\; I_{o}l\; \sin \; \theta}{4\pi \; r}\left\lbrack {1 + \frac{1}{j\; {kr}} - \frac{1}{({kr})^{2}}} \right\rbrack} \cdot ^{{- j}\; {kr}}}}} & {{Equation}\mspace{20mu} 13} \\{H_{\varphi} = {j{{\frac{k\; I_{o}l\; \sin \; \theta}{4\pi \; r}\left\lbrack {1 + \frac{1}{j\; {kr}}} \right\rbrack} \cdot ^{{- j}\; {kr}}}}} & {{Equation}\mspace{20mu} 14}\end{matrix}$

In these equations, r is the distance to the antenna and not the antennaradius. After some algebraic manipulations, the following simplifiedequations for the field magnitude can be obtained:

$\begin{matrix}{{E_{\theta}}^{2} \propto {\frac{1}{({kr})^{2}} - \frac{1}{({kr})^{4}} + \frac{1}{({kr})^{6}}} \propto P_{{RX},E}} & {{Equation}\mspace{20mu} 15} \\{{H_{\varphi}}^{2} \propto {\frac{1}{({kr})^{2}} + \frac{1}{({kr})^{4}}} \propto P_{{RX},H}} & {{Equation}\mspace{20mu} 16}\end{matrix}$

The received power from a co-polarized antenna, that is, one in whichthe transmit and the receive antenna are parallel to each other, isproportional to the time averaged value of the incident field squared asdescribed above. Thus, the path gain can be calculated as follows

$\begin{matrix}{G_{{path},E_{\varphi}} = {\frac{P_{{RX},E}}{P_{TX}} = {\frac{G_{Tx}G_{Rx}}{4}\left\lbrack {\frac{1}{({kr})^{2}} - \frac{1}{({kr})^{4}} + \frac{1}{({kr})^{6}}} \right\rbrack}}} & {{Equation}\mspace{20mu} 17} \\{G_{{path},H_{\varphi}} = {\frac{P_{{RX},H}}{P_{TX}} = {\frac{G_{Tx}G_{Rx}}{4}\left\lbrack {\frac{1}{({kr})^{2}} + \frac{1}{({kr})^{4}}} \right\rbrack}}} & {{Equation}\mspace{20mu} 18}\end{matrix}$

Equation 17 is the propagation law for like antennas (propagation from adipole to another co-polarized dipole or propagation from a loop toanother co-polarized loop). Equation 18 is the propagation law forunlike antennas (propagation from a dipole to a co-polarized loop orpropagation from a loop to a co-polarized dipole). The path gain in thenear-field is much higher than what would be expected by applying thefar-field theory (Friis equation). For the transmission between likeantennas in the near-field a path loss of 60 dB/decade can be seen,whereas the transmission between unlike antennas in the near-field has apath loss of 40 dB/decade. This is contrasted to the path loss of 20dB/decade that is seen in the far-field.

These equations can be used to determine additional antennas andcharacteristics that can be used for this purpose.

Although only a few embodiments have been disclosed in detail above,other embodiments are possible and the inventors intend these to beencompassed within this specification. The specification describesspecific examples to accomplish a more general goal that may beaccomplished in another way. This disclosure is intended to beexemplary, and the claims are intended to cover any modification oralternative which might be predictable to a person having ordinary skillin the art.

For example, other antenna forms and selections can be used. The term“power” as used herein can refer to any kind of energy, power or forcetransfer of any type. The receiving source can be any device thatoperates from stored energy, including a computer or peripheral,communicator, automobile, or any other device. Myriad applications ofthe foregoing transmitter, receiver and transceiver apparatus of theinvention are recognized. By way of example and without limitation, suchapplications include: (i) powering or charging portable computers. PMDs.client devices. cellular phones. etc.; (ii) powering or charging flatscreen or wall-mounted televisions or displays; (iii) powering orcharging refrigerators (e.g., by placing a transmitter on the wallbehind the refrigerator. and a receiver in the refrigerator proximate tothe transmitter); (iv) powering or charging electric cars; e.g., byplacing or building in a transmitter in the floor of a garage, andplacing a receiver on the bottom of the car: (v) powering or charginghome or office lighting: e.g. incandescent, fluorescent or LED-basedlamps with no cords: (vi) powering or charging home or office appliancessuch as toasters, blenders, clocks, televisions, microwave ovens,printers, computers, etc.: (vii) powering or charging multiple devicessimultaneously (e.g., through the use of a substantiallyomni-directional transmitter arrangement); and (viii) powering orcharging devices where the presence of electrical conductors withvoltage would represent a hazard; e.g., near water, near children, etc.

As used herein, the terms “electrical component” and “electroniccomponent” are used interchangeably and refer to, without limitation,components adapted to provide some electrical or electronic function,including without limitation inductive reactors (“choke coils”),transformers, filters, gapped core toroids, inductors, capacitors,resistors, operational amplifiers, varactors, MEMS devices, FETs andother transistors and diodes, whether discrete components or integratedcircuits, whether alone or in combination.

As used herein, the term “integrated circuit (IC)” refers to any type ofdevice having any level of integration (including without limitationULSI, VLSI, and LSI) and irrespective of process or base materials(including, without limitation Si, SiGe, CMOS and GAs). ICs may include,for example, memory devices (e.g., DRAM, SRAM, DDRAM, EEPROM/Flash,ROM), digital processors, SoC devices, FPGAs, ASICs, ADCs, DACs andother devices, as well as any combinations thereof.

As used herein, the term “digital processor” is meant generally toinclude all types of digital processing devices including, withoutlimitation, digital signal processors (DSPs), reduced instruction setcomputers (RISC), general-purpose (CISC) processors, microprocessors,gate arrays (e.g., FPGAs), Reconfigurable Compute Fabrics (RCFs), andapplication-specific integrated circuits (ASICs). Such digitalprocessors may be contained on a single unitary IC die, or distributedacross multiple components.

As used herein, the terms “computing device”, “client device”, and “enduser device” include, but are not limited to, personal computers (PCs)and minicomputers, whether desktop, laptop, or otherwise, set-top boxessuch as the Motorola DCT2XXX15XXX and Scientific Atlanta Explorer2XXX13XXX/4XXX18XXX series digital devices, personal digital assistants(PDAs) such as the Blackberry® or “Palm®” family of devices, handheldcomputers, personal communicators, J2ME equipped devices, cellulartelephones, or literally any other device capable of using power, orinterchanging data with a network.

As used herein, the term “memory” includes any type of integratedcircuit or other storage device adapted for storing digital dataincluding, without limitation, ROM, PROM, EEPROM, DRAM, SDRAM, DDR/2SDRAM, EDOIFPMS, RLDRAM, SRAM, “flash” memory (e.g., NANDINOR), andPSRAM.

Also, the inventors intend that only those claims which use the words“means for” are intended to be interpreted under 35 USC 112, sixthparagraph. Moreover, no limitations from the specification are intendedto be read into any claims, unless those limitations are expresslyincluded in the claims.

The operations and/or flowcharts described herein may be carried out ona computer, or manually. If carried out on a computer, the computer maybe any kind of computer, either general purpose, or some specificpurpose computer such as a workstation. The computer may be an Intel(e.g., Pentium or Core 2 duo) or AMD based computer, running Windows XPor Linux, or may be a Macintosh computer. The computer may also be ahandheld computer, such as a PDA, cellphone, or laptop.

Moreover, the method steps and operations described herein can becarried out on a dedicated machine that does these functions.

The programs may be written in C or Python, or Java, Brew or any otherprogramming language. The programs may be resident on a storage medium,e.g., magnetic or optical, e.g. the computer hard drive, a removabledisk or media such as a memory stick or SD media, wired or wirelessnetwork based or Bluetooth based Network Attached Storage (NAS), orother removable medium or other removable medium. The programs may alsobe run over a network, for example, with a server or other machinesending signals to the local machine, which allows the local machine tocarry out the operations described herein.

Where a specific numerical value is mentioned herein, it should beconsidered that the value may be increased or decreased by 20%, whilestill staying within the teachings of the present application, unlesssome different range is specifically mentioned. Where a specifiedlogical sense is used, the opposite logical sense is also intended to beencompassed.

1. A method, comprising: inducing power into a near field magnetic fieldof a transmitting antenna; and operating the transmitting antenna alongwith a receiving antenna, to couple the power magnetically to areceiving antenna that has at least one characteristic that is matchedto the transmitting antenna; and tuning at least one of said antennas toimprove a matching therebetween.
 2. A method as in claim 1, wherein saidmatching comprises matching impedances between said antennas.
 3. Amethod as in claim 2, wherein said tuning comprises using an adjustablecapacitor in the at least one of said antennas and electricallyadjusting a capacitance value of the adjustable capacitor to a desiredvalue.
 4. A method as in claim 3, wherein said tuning comprisesadjusting a distance between plates in a capacitor.
 5. A method as inclaim 4, wherein said adjusting comprises using screws made of anon-electrical material.
 6. A method as in claim 3, wherein saidadjustable capacitor comprises an electrically adjustable capacitor, andsaid tuning comprises automatically tuning.
 7. A method as in claim 6,wherein said electrically adjustable capacitor comprises a varactordiode.
 8. A method as in claim 3, further comprising a fixed valuecapacitor in addition to said adjustable capacitor.
 9. A method as inclaim 8, wherein said variable capacitor has 10% of a capacitance valueof the fixed value capacitor.
 10. A method as in claim 2, furthercomprising detecting a temporary situation that reduces matching betweenthe transmitter and receiver, and carrying out at least one operationresponsive to said detecting.
 11. A method as in claim 10, wherein saiddetecting is carried out at the transmitter.
 12. A method as in claim11, wherein said least one operation comprises terminating powertransmission until the temporary situation is terminated.
 13. A methodas in claim 2 wherein said tuning comprises using an electricallyadjustable capacitor in the at least one antenna and electricallyadjusting a capacitance value of the electrically-adjustable capacitorto a desired value and wherein said at least one operation compriseschanging a value of said adjustable capacitor.
 14. A method as in claim1, wherein said transmitter and said receiver are both formed ofantennas made of circular shaped loops of electrically conductivematerial.
 15. A method as in claim 1, wherein said operating comprisesstoring energy in and near field of the transmitting antenna andinducing said energy into the receiving antenna.
 16. A method as inclaim 1, wherein each of said antennas have a Q of at least
 1000. 17. Amethod as in claim 1, wherein each of said antennas are magneticantennas which are tuned to within 10% of their resonant values.
 18. Amethod as in claim 2, wherein said antennas include capacitors therein,and further comprising sizing said capacitors to withstand at least 2 KVof reactive voltage.
 19. A wireless power transmitter, comprising: apower transmitter, that produces power to be transmitted wirelessly overa magnetic link; and an inductive loop antenna part, having at least onecapacitive part connected to said inductive loop part, wherein saidcapacitive part is sized to withstand at least 2 KV of reactive voltage.20. A transmitter as in claim 19, further comprising a receiver with areceiving antenna that is matched to the transmitting antenna.
 21. Atransmitter as in claim 19, further comprising a tuning part that allowstuning said antenna.
 22. A transmitter as in claim 21, wherein saidtuning part comprises an adjustable capacitor.
 23. A transmitter as inclaim 22, wherein said adjustable capacitor comprises a capacitor withplates whose distance can be adjusted.
 24. A transmitter as in claim 23,wherein said capacitor has screws made of a non-electrical material. 25.A transmitter as in claim 21, wherein said tuning part comprises anelectrically adjustable capacitor.
 26. A transmitter as in claim 25,wherein said tuning part comprises a varactor diode.
 27. A transmitteras in claim 25, further comprising a fixed value capacitor in additionto said adjustable capacitor.
 28. A transmitter as in claim 27, whereinsaid variable capacitor that has 10% of a capacitance value of the fixedvalue capacitor.
 29. A transmitter as in claim 20, further comprising apart that detects a reduction of matching between the transmitter andreceiver, and carries out at least one operation responsive to thedetecting.
 30. A transmitter as in claim 29, wherein said deviceterminates power transmission until the temporary situation isterminated.
 31. A transmitter as in claim 29, further comprising anadjustable capacitor in at least one antenna and said deviceelectrically adjusts a capacitance value of the electrically-adjustablecapacitor to a desired value to change a value of said adjustablecapacitor to improve said matching.
 32. A transmitter as in claim 19,wherein said antenna is made of circular shaped loop of electricallyconductive material.
 33. A transmitter as in claim 19, wherein saidantenna has a Q of at least
 1000. 34. A transmitter as in claim 19,wherein said antenna is a magnetic antenna which is tuned to within 10%of its resonant value.
 35. A transmitter as in claim 19, wherein saidantenna has a first part, connected to the power transmitter, andforming a coupling loop, and a second part, electrically unconnected tosaid coupling loop, into which power is electrically induced.
 36. Atransmitter as in claim 19, wherein at least part of said antenna iselectrically decoupled from said power transmitter.
 37. A transmitter asin claim 19, wherein at least part of said antenna is coated with ananti corrosion material.
 38. A wireless power receiver, comprising: aninductive loop antenna part, having at least one capacitive partconnected to said inductive loop part, wherein said capacitive part issized to withstand at least 2 KV of reactive voltage; and a powerreceiver, coupled to receive power from said inductive loop antennapart, that has been transmitted wirelessly over a magnetic link.
 39. Areceiver as in claim 38, wherein said inductive loop antenna part formsa receiving antenna, and further comprising a transmitter with atransmitting antenna that is matched to the receiving antenna.
 40. Areceiver as in claim 38, further comprising a tuning part that allowstuning said antenna.
 41. A receiver as in claim 40, wherein said tuningpart comprises an adjustable capacitor.
 42. A receiver as in claim 41,wherein said adjustable capacitor comprises a capacitor with plateswhose distance can be adjusted.
 43. A receiver as in claim 42, whereinsaid capacitor has screws made of a non-electrical material.
 44. Areceiver as in claim 40, wherein said tuning part comprises anelectrically adjustable capacitor.
 45. A receiver as in claim 44,wherein said tuning part comprises a varactor diode.
 46. A receiver asin claim 44, further comprising a fixed value capacitor in addition tosaid adjustable capacitor.
 47. A receiver as in claim 46, wherein saidvariable capacitor that has 10% of a capacitance value of the fixedvalue capacitor.
 48. A receiver as in claim 39, further comprising apart that detects a reduction of matching between the receiver andtransmitter, and carries out at least one operation responsive to thedetecting.
 49. A receiver as in claim 48, wherein said device terminatespower transmission until the temporary situation is terminated.
 50. Areceiver as in claim 48, further comprising an adjustable capacitor inat least one antenna and said device electrically adjusts a capacitancevalue of the electrically-adjustable capacitor to a desired value tochange a value of said adjustable capacitor to improve said matching.52. A receiver as in claim 38, wherein said antenna is made of acircular shaped loop of electrically conductive material.
 53. A receiveras in claim 38, wherein said antenna has a Q of at least
 1000. 54. Areceiver as in claim 38, wherein said antenna is a magnetic antennawhich is tuned to within 10% of its resonant value.
 55. A receiver as inclaim 38, wherein at least part of said antenna is coated with an anticorrosion material.
 56. A wireless power transmitter, comprising: apower transmitter connection, that receives energy to be transmittedwirelessly over a magnetic link; a first coupling antenna part,electrically connected to receive said energy; and a second antennapart, electrically disconnected from said first coupling part and fromsaid power transmitter connection, and operating to transmit,magnetically, said energy.
 57. A transmitter as in claim 56, wherein atleast one of said antenna parts includes an inductive loop antenna part,having at least one capacitive part connected to said inductive looppart, wherein said capacitive part is sized to withstand at least 2 KVof reactive voltage.
 58. A transmitter as in claim 56, furthercomprising a tuning part that allows tuning said antenna.
 59. Atransmitter as in claim 58, wherein said tuning part comprises anadjustable capacitor.
 60. A transmitter as in claim 56, wherein saidadjustable capacitor comprises a capacitor with plates whose distancecan be adjusted.
 61. A transmitter as in claim 58, wherein said tuningpart comprises an electrically adjustable capacitor.
 62. A transmitteras in claim 61, wherein said tuning part comprises a varactor diode. 63.A transmitter as in claim 61, further comprising a fixed value capacitorin addition to said adjustable capacitor.
 64. A transmitter as in claim63, wherein said variable capacitor that has 10% of a capacitance valueof the fixed value capacitor.
 65. A method, comprising: transmittingpower from a transmitting antenna to a receiving antenna; automaticallydetecting a detuning event that detunes a relationship between saidtransmitting antenna and receiving antenna; and responsive to saiddetecting, automatically taking an action to change a characteristic ofsaid transmitting.
 66. A method as in claim 65, wherein said actioncomprises terminating device power transmission during a time of thedetuning event.
 67. A transmitter as in claim 65, wherein said actioncomprises electronically tuning one of said antennas to a differentresonance value.
 68. A wireless power transmitter, comprising: a powertransmitter connection, that receives energy to be transmittedwirelessly over a magnetic link; at least one antenna part formed of asingle loop of conductive material; and a capacitor part, having a valueto match an L and C value of the antenna to a frequency of said powertransmitter.
 69. A transmitter as in claim 68, wherein said antennaincludes a first coupling antenna part, electrically connected toreceive said energy; and a second antenna part, electricallydisconnected from said first coupling part and from said powertransmitter connection, and operating to transmit, magnetically, saidenergy.
 70. A transmitter as in claim 68, wherein said capacitor part issized to withstand 2 KV of reactive voltage.
 71. A method, comprising:magnetically transmitting power from a transmitting antenna to areceiving antenna, where each of said antennas has a Q value for aspecified frequency, greater than 1000; and tuning said antennas tomaintain resonance at said specified frequency.
 72. A method comprising:forming a radiated magnetic field with energy therein using atransmitting antenna with a Q value greater than 1000; extracting energyfrom the radiated field of the transmitting antenna, at a receivingantenna location that is unconnected to said transmitting antenna by anywires; and using said energy in an electronic device that is unconnectedto said transmitting antenna by any wires.
 73. A method as in claim 72,further comprising adaptively tuning circuit one of said antennas.
 74. Amethod as in claim 73 wherein said adaptive tuning is carried outautomatically by an electric circuit associated with at least one ofsaid antennas.
 75. A method as in claim 72, further comprising matchinga resistance of a receiving antenna to the load resistance.
 76. A methodas in claim 72, further comprising shielding at least one of saidantennas against external influences to resonance.
 77. A method as inclaim 76, wherein said shield is against stray capacitance.
 78. Anapparatus comprising: a receiving antenna, operative to extract energyfrom a radiated field at a receiving antenna location that isunconnected to any transmitting antenna by any wires; and a powersystem, using said energy in an electronic device that is unconnected tosaid transmitting antenna by any wires.
 79. An apparatus as in claim 78,further comprising an adaptive tuning circuit, changing a tuning valueof one of said antennas.
 80. An apparatus as in claim 79 wherein saidadaptive tuning circuit is an automatic tuning circuit associated withat least one of said antennas.
 81. An apparatus as in claim 78, furthercomprising a matching circuit that matches a resistance of a receivingantenna to the load resistance.
 82. An apparatus as in claim 78, furthercomprising a shield that shields at least one of said antennas againstexternal influences to resonance.
 83. An apparatus as in claim 82,wherein said shield is a slotted shield.